Compact, Dual Band Antenna for Sub-GHz Internet-of-Things (IoT) Applications

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Compact, Dual Band Antenna for Sub-GHz

Internet-of-Things (IoT) Applications

Frederic Anthierens

Student number: 01407926

Supervisors: Prof. dr. ir. Hendrik Rogier, Prof. dr. ing. Patrick Van Torre

Counsellor: ing. Thomas Ameloot

Master’s dissertation submitted in order to obtain the academic degree of

Master of Science in Electrical Engineering - main subject Communication and Information Technology

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Compact, Dual Band Antenna for Sub-GHz

Internet-of-Things (IoT) Applications

Frederic Anthierens

Supervisors: Prof. dr. ir. Hendrik Rogier, Prof. dr. ing. Patrick Van Torre

Counsellor: ing. Thomas Ameloot

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Admission to loan

The author gives permission to make this master dissertation available for consultation and to copy parts of this master dissertation for personal use. In all cases of other use, the copyright terms have to be respected, in particular with regard to the obligation to state explicitly the source when quoting results from this master dissertation.

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Preamble

The radiation pattern of the eighth-mode cavity antenna and the dual band antenna could both not be measured due to the COVID-19 epidemic (2020). Instead, both measurement campaigns are replaced with the simulation of an Artificial Magnetic Conductor (AMC) plane. The simulation is carried out in free-space and in proximity of the human body.

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Preface

In this thesis, the design of a textile antenna in the sub-GHz frequency band is investigated. I find it an interesting and challenging research topic. As the year has passed, it was a pleasure to delve into this subject and to solve the challenges that I have encountered along the way. The obtained results would not have been possible without the input of my two supervisors Prof. dr. ir. Hendrik Rogier, Prof. dr. ing. Patrick Van Torre and my counsellor ing. Thomas Ameloot. Therefore, I would like to thank them for their thoughtful guidance and their helpful advice throughout my thesis.

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Compact, Dual Band Antenna for Sub-GHz

Internet-of-Things (IoT) Applications

Frederic Anthierens

Supervisors: Prof. dr. ir. Hendrik Rogier, Prof. dr. ing. Patrick Van Torre Counsellor: ing. Thomas Ameloot

Academic year 2019-2020 Ghent University

Abstract

A wearable textile dual band antenna based on Substrate Integrated Waveguide (SIW) tech-nology that operates in the 434 MHz and the 868 MHz Industrial, Scientific and Medical (ISM) bands, is proposed. A monopole antenna is designed for the 434 MHz ISM band and the 868 MHz ISM band is covered by an eighth-mode cavity antenna. The design methodology relies on the meander technique and the virtual magnetic wall technique to reduce the dimensions of the monopole and the cavity antenna, respectively. Unfortunately, the proximity of the human body has a considerable impact on the performance of the monopole antenna. An Ar-tificial Magnetic Conductor (AMC) plane is introduced to increase the front-to-back ratio and to isolate the dual band antenna from the human body. Based on the investigation of different unit cells, the interwoven spiral unit cell is used since its small dimensions lead to a portable structure (dual band antenna and AMC plane). The dimensions of the structure are optimized in free-space to achieve a high front-to-back ratio and a good reflection coefficient within the desired frequency bands. Finally, the performance of the structure is simulated in proximity of the human body.

Keywords

Textile dual band antenna, Substrate Integrated Waveguide (SIW), Meander monopole an-tenna, Eighth-mode Substrate Integrated Waveguide (EMSIW), Artificial Magnetic Conductor (AMC)

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Counsellor: ing. Thomas Ameloot

Abstract—A wearable textile dual band antenna based on substrate integrated waveguide technology that operates in the 434 MHz and the 868 MHz Industrial, Scientific and Medical (ISM) bands, is proposed. A monopole antenna is designed for the 434 MHz ISM band and the 868 MHz ISM band is covered by an eighth-mode cavity antenna. The design methodology relies on the meander technique and the virtual magnetic wall technique to reduce the dimensions of the monopole and the cavity antenna, respectively. In addition to the design of the dual band antenna, an AMC plane is proposed to increase the front-to-back ratio and to isolate the dual band antenna from the human body. The dimensions of the structure (dual band antenna and AMC plane) are optimized in free-space to achieve a high front-to-back ratio and a good reflection coefficient within the desired frequency bands. Finally, the performance of the structure is simulated in proximity of the human body.

Index Terms—Textile dual band antenna, GHz, Sub-strate Integrated Waveguide (SIW), Meander monopole antenna, Eighth-mode Substrate Integrated Waveguide (EMSIW), Artifi-cial Magnetic Conductor (AMC)

I. INTRODUCTION

T

HE Internet of Things (IoT) is a rapidly evolving research field and its applications form the research topic for a variety of studies. The Internet of Things consists of a number of smart devices that are connected to a network and are able to communicate with each other. Several networks are possible: mobile networks, LoRa-networks, Sigfox-networks, Ethernet-connections. . . The LoRa-network and Sigfox-networks are specifically designed for IoT-applications. The smart devices contain several sensors to collect information about their en-vironment and forward it over the network. At a central place, the information is collected and analyzed. It is expected by Proximus that 150 billion devices worldwide will be connected to the IoT by 2025 [1]. The number of smart devices will further increase and fulfill a more prominent role in our everyday life. Smart devices try to make our life easier and more comfortable. Their applications are uncountable and vary from a smart shower head that can play your favorite music to smart textiles. These textiles are able to integrate several electronic components such as sensors to detect body and environmental parameters or an actuator that can give a signal to the wearer. These textiles also contain a communication device that establishes a wireless communication link via an antenna to a nearby base station. The antenna could be man-ufactured with textile materials. These textile antennas have specific requirements and the proximity of the human body makes the design more challenging. The ease of integration

into the garment of the wearer is the most important design requirement. A bulky, heavy antenna will not be appealing to potential customers and therefore will not be a commercial success. The ease of integration is mainly determined by the size of the antenna. Unfortunately, the size reduction comes at a prize: smaller antennas lead to a poorer performance. Besides the dimensions of the antenna, the ease of integration is also determined by the weight, the flexibility and the complexity of the feeding structure of the antenna. A second important design criterium is that the antenna must be isolated from the human body. The bandwidth should be large enough to compensate the frequency shifts, caused by the deformations of the antenna during its use or by the proximity of the human body. The proximity of the human body can also cause a deformation of the radiation pattern. Therefore, it is important that the antenna has a high front-to-back ratio. The antenna will then enable more robust and energy efficient wireless communication. Additionally, less energy will be dissipated into the human body and the likelihood of exceeding the Specific Absorption Rate (SAR) limits will be smaller.

A lot of research has been conducted on textile antennas. The current academic literature is mainly focused on textile antennas that operate in the GHz bands and less in the sub-GHz bands. The development of the Substrate Integrated Waveguide (SIW) technology paved the way for the implemen-tation of rectangular waveguides to create high-performance textile antennas and other microwave components in planar form [2]. These waveguide structures contain two parallel conducting planes with in between a dielectric substrate. The two parallel planes are connected with two rows of conducting cylinders or eyelets. The fabrication process can become very complicated due to a large number of eyelets (puncturing steps) as well as embroidery steps. Researchers have tried to decrease the number of fabrication steps, in order to make the SIW technology more appealing for the industry. The metal eyelets can be omitted by cutting the top and bottom metal layers in one piece from a single e-textile sheet and fold them around the substrate [3]. The embroidery steps can then be avoided by making use of an adhesive sheet to connect the metal layers to the substrate. This fabrication method leads to a more light-weight, compact and mechanically more flexible antenna. The robustness of the antenna is increased because tearing of the electrotextile around the metal eyelets has now been avoided. Miniaturization techniques that exploit the shape of the antenna to reduce its dimensions, offer attractive opportunities to the designer. The meander technique tries to

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2 fill the space more efficiently by bending a long straight line

to occupy an overall shorter length [4]. However, the fields from different parts of a strip line can cancel each other this way and care should be taken to minimize the losses. Another miniaturization approach makes advantage of the symmetry present in the mode profile of the magnetic field. A half-mode structure is obtained by bisecting the cavity antenna along the symmetry plane [5], [6]. The technique can be repeated to create quarter-mode [7], eighth-mode [8], 16th-mode [9] and even 64th-mode antennas [10].

Introducing an Artificial Magnetic Conductor (AMC) plane to the design, could increase the front-to-back ratio and isolate the antenna from the human body. The reflection coefficient of the AMC plane is +1 at its resonance frequency. The AMC plane can be closely located to a parallel current on the antenna forming a reflector, since the parallel current is in phase with its image on the AMC plane. The bandwidth of the AMC plane is defined by the phase of the reflecting wave, lying in the interval 0◦ to ±90◦ [11]. If only a single resonance frequency is required, the square patch unit cell can be used to design the AMC plane [11], [12]. The simple shape eases the fabrication and decreases the risk for manufacturing inaccuracies. Moreover, the square patch unit cell has a broad bandwidth. The dimensions of the unit cell can be reduced by incorporating a slot into the center of the square patch [13], [14], [15]. Another attempt to reduce the unit cell size is based on space filling curves [16], [17]. The unit cells of paper [16] and [17] consist of two dipoles, which intersect each other perpendicularly at the center. The arms of the dipole are meandered. Unfortunately, the unit cell suffers from a narrow bandwidth, but can be increased by interleaving the arms of the dipole to adjacent unit cells. This approach not only leads to a larger bandwidth, but also to a further reduction of the dimensions of the unit cell.

This manuscript outlines the design of a dual band SIW antenna, which is intended for an application in the 434 MHz and 868 MHz ISM bands such as those used for LoRa communication. A bandwidth of 20 MHz at each resonance frequency (424 MHz to 444 MHz and 858 MHz to 878 MHz) should be sufficient [18]. The dual band antenna is aimed to be integrated in the garment of rescue workers to provide the communication between the rescue worker and the base station. The design process of the dual band antenna is explained in section II. The design of the AMC plane and overall structure (dual band antenna with the AMC plane) are outlined in section III and section IV.

II. DUAL BAND ANTENNA DESIGN

A. Antenna topology

This section proposes a dual band SIW antenna operating in the 434 MHz and in the 868 MHz ISM bands. The dual band antenna is excited with a low complexity probe feed. The topology of the designed dual band antenna is depicted in Fig. 1. The dimensions of the antenna are equal to 140 mm x 140 mm x 8 mm. In order to obtain an antenna with a good radiation efficiency, it is important to use a substrate with low losses and an e-textile with a low sheet resistance. The

substrate is realized with a closed-cell expanding rubber (r = 1.32, tanδ = 0.015) and the metal layers are implemented with a copper-plated taffeta polyester (σ = 3 · 105_S/m).

Fig. 1: The topology of the dual band antenna, excited by a low-complexity probe feed

1) Eighth-mode cavity antenna

The eighth-mode cavity antenna covers the 868 MHz ISM band. The design of the cavity antenna is based on the the virtual magnetic wall technique. The design procedure is described in [8]. The virtual magnetic wall technique is a miniaturization technique based on the mode profile of the magnetic field. The tangential component of the magnetic field vanishes alongside the symmetry planes of the cavity, so they act like a Perfect Magnetic Conductor (PMC). A half-mode structure is obtained by bisecting the cavity antenna along the symmetry plane. The new structure has a quasi-identical mode profile and operating frequency but only occupies half of the space of the original structure. An eighth-mode cavity antenna is constructed by applying this technique three times. The procedure from [8] enhances the bandwidth by placing two such resonators with slightly different resonance frequencies in close proximity on a common ground plane. The fields of both cavities will couple and only one cavity (main resonator) requires a feeding network while the other cavity acts as a coupled resonator. The coupled resonator could not be re-scaled to the 434 MHz ISM band. The dimensions would become too large and the antenna structure would no longer be portable. Therefore, the coupled resonator is replaced by a monopole to keep the dimensions of the antenna structure limited.

2) Monopole antenna

The monopole is placed alongside the edge of the cavity. The beginning of the monopole has the largest contribution to the far field because this is the place where the magnitude of the current is the biggest. Therefore, it is important that the monopole is placed directly above the zone where the ground plane is removed. The ground plane is displayed in Fig. 2 and will be discussed further on in this section. The meander technique [4] is applied at the end of the monopole, where the

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deterioration of the performance. The width of the monopole is set to 5 mm. The monopole will be cut out by hand from the e-textile. If the width of the monopole is too narrow, it is expected that the conducting textile will unravel and too little threads will be available to conduct the current. Laser cutting in combination with a proper coating material might be able to counter the unraveling problem. The starting point of the monopole has been broadened to 36 mm. This offers multiple current paths for a λ/4 resonance peak and leads to an enhancement of the bandwidth.

3) Ground plane

The ground plane of the dual band antenna is shown in Fig. 2. This ground plane is extended beyond the radiating apertures of the eighth-mode cavity antenna to obtain a more directive radiation pattern [8]. Ideally, the size of the ground plane is very large to have a directive radiation pattern. This leads to a high isolation between the antenna and the human body and avoids resonance frequency detuning or radiation pattern deformation. Unfortunately this is not possible because this hinders an easy integration of the dual band antenna into the garment of the wearer. The size of the ground plane is kept to a minimum because the ease of integration is the primary design requirement. Furthermore, the ground plane underneath the monopole must be removed. Otherwise, the current on the monopole would cancel out with its image on the ground plane and the performance of the monopole will be deteriorated. The ground plane in the upper left corner is removed to further decrease the capacitive coupling between the monopole and the ground plane. It improves the performance of the monopole while leaving the performance of the cavity almost untouched.

Fig. 2: The ground plane of the dual band antenna 4) Low-complexity probe feed

The dual band antenna is excited by a low-complexity probe feed. The position of the feed is marked with a small black dot in Fig. 1. The monopole is matched to the cavity antenna

radiated or converted into heat due to the presence of losses. The antenna is intended for a LoRa-application, therefore a bandwidth of 20 MHz at each resonance frequency should be sufficient (424 MHz - 444 MHz and 858 MHz - 878 MHz) [18]. Fig. 3 shows that the condition in the two operational bands (yellow rectangle) is fulfilled. The resonance frequency of the monopole is equal to 432.4 MHz and the resonance frequency of the cavity antenna is equal to 869.2 MHz. The monopole has a bandwidth equal to 28.5 MHz and the bandwidth of the cavity is equal to 29 MHz.

Fig. 3: The magnitude of the simulated reflection coefficient of the dual band antenna in free-space 6) Free-space antenna gain

The simulated free-space antenna gain at 434 MHz is depicted in Fig. 4(a). The main beam of the pattern is directed towards the user. The other beam is directed away from the user and has a magnitude equal to 0.68 dBi. The simulated free-space antenna gain at 868 MHz is given in Fig. 4(b). The pattern is much more directive in comparison to that of the monopole. The magnitude of the main lobe is equal to 1.46 dBi and has a beam width equal to 146◦.

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Fig. 4: The simulated free-space antenna gain [dBi] in the vertical (xz) plane at 434 MHz (a) and at 868 MHz (b)

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4 B. Influence of the human body

The performance of the antenna is affected by the prox-imity of the human body, which can cause a detuning of the resonance frequency or a deformation of the radiation pattern [8]. The simulation becomes very complex when all the different influences of the human body on the antenna are taken into consideration. Therefore, it is assumed that the antenna is not bent or deformed during its use. The human body is simulated with a multilayered tissue model [19]. The cross-section of the model is presented in Fig. 5 and contains 6 layers. The electrical parameters of each human tissue are frequency-dependent and can be found in [20], [21] and [22].

Fig. 5: The cross-section of the body model

The influence of the human body on the performance of the dual band antenna is investigated by varying the gap size between the antenna and the textile layer (r = 2.231, σ = 0.0366 S/m) of the body model from 0 mm to 24 mm. The magnitude of the reflection coefficient for the different gap sizes is visualized in Fig. 6. The resonance peak of the monopole antenna shifts to lower frequencies when the an-tenna is placed closer to the body model, while the resonance peak of the eighth-mode cavity antenna remains relatively unaltered. This different behavior of both antennas can be explained by their gain patterns in Fig. 4. The gain pattern of the monopole is less directional in comparison to the gain pattern of the eighth-mode cavity antenna. A large part of the fields of the monopole are radiated towards the human body such that the proximity of the human body will have a larger impact on the performance of the monopole antenna.

Fig. 6: The magnitude of the simulated reflection coefficient for a gap size from 0 mm to 24 mm

III. AMCPLANE DESIGN

The proximity of the human body greatly decreases the performance of the monopole. The interaction between the antenna and the human body is decreased by placing a spacer between them. The spacer is realized with a plastic air cushion. The electrical parameters of the plastic air cushion are similar to air (r = 1.00059). Fig. 6 (see previous section) shows that the dual band antenna meets his design requirements if the distance between the dual band antenna and the human body is greater than 24 mm. The introduction of the spacer would lead to a significant increase in the thickness of the overall structure. In addition, the AMC plane with the zero crossing of the phase placed at 434 MHz, is introduced to isolate the dual band antenna from the human body. The AMC plane increases the front-to-back ratio and limits the thickness of the spacer. The AMC plane is designed based on the interwoven spiral unit cell and is simulated in CST Microwave Studio [16]. The infinite planar periodicity is mimicked by setting the boundary conditions to unit cell with the corresponding Floquet ports. The substrate of the AMC plane is implemented with a closed-cell expanding rubber (r= 1.32, tanδ = 0.015) and the metal layers are realized with copper-plated taffeta polyester (σ = 3·105_{S/m). The plastic air cushion spacer between the antenna} is included in the simulation of the unit cell. The thickness of the spacer is 15 mm and should be just enough to integrate the SMA cable to the feed of the dual band antenna. The topology and cross-section of the spiral unit cell are shown in Fig. 7. The unit cell consists of two dipoles, which intersect each other perpendicularly at the center. The arms of the dipoles are meandered clockwise around the center of the unit cell.

Fig. 7: Topology and cross-section of the interwoven spiral unit cell

The basic behavior around the fundamental resonance fre-quency can be described fairly accurately by a LC model [16]. The model consists of an inductance L with a series capacitance C. The inductance L is mainly determined by the length of the spiral conductor and the capacitance C is caused by the capacitive coupling between the adjacent turns. The resonance frequency (fr ∼ 1/√LC) and the

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and a decrease in dimensions. Fig. 8 shows the magnitude and phase of the simulated reflection coefficient with a plane wave incident from the normal direction. The reference plane is placed on top of the spacer. The phase of the reflection coefficient has a zero crossing at 434 MHz. The bandwidth is small because the phase exhibits a strong descending slope around the resonance frequency.

Fig. 8: Simulated reflection coefficient of the interwoven spiral unit cell

The AMC plane is formed by 3x3 interwoven spiral unit cells. The topology and cross-section are shown in Fig. 9. A ground plane has been added to the AMC plane, in order to obtain a single resonance peak around 434 MHz. The thickness of the substrate is increased from 4 mm to 8 mm to preserve the performance of the spiral resonator.

Fig. 9: The topology of the AMC surface

reflection coefficient below -10 dB at 434 MHz. A 30 mm thick lossy material is placed underneath the structure (dual band antenna and AMC plane) to check how well the AMC plane can isolate the dual band antenna from the human body. The electrical parameters of the lossy material are set to the parameters of human skin and are found in [20], [21] and [22]. The magnitude of the simulated reflection coefficient of the structure, in proximity of the lossy material, is visualized in Fig. 10. The gap size between the structure and the human skin is increased from 0 mm to 30 mm. The figure shows that the magnitude remains unchanged when the gap size increases. Therefore, it can be stated that the AMC plane successfully shields the dual band antenna from the human body and that no additional spacer is needed between the AMC plane and the human body.

Fig. 10: The influence of the gap size on the magnitude of the simulated reflection coefficient of the structure in

proximity of a 30 mm thick layer of human skin The human body is simulated with the multilayered tissue model from section 3. The magnitude and phase of the simulated reflection coefficient of the structure in proximity of the human body is displayed in Fig. 11. The magnitude of the simulated reflection coefficient exhibits approximately the same curve as the simulated reflection coefficient in Fig. 10. This shows that the AMC plane largely isolates the antenna from the human body and that the human body has only a small influence on the reflection coefficient of the dual band antenna. The bandwidth of the monopole and the eighth-mode cavity antenna are respectively equal to 36 MHz and to 35.7 MHz. Furthermore, the structure is able to cover both bands of operation.

The simulated antenna gain at 434 MHz is given in Fig. 12(a). Compared to the gain pattern in Fig. 4(a), the front-to-back ratio has drastically improved despite the proximity of the human body. The main lobe is now directed away from the user and its magnitude is increased to -9.36 dBi. The simulated

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6 antenna gain at 868 MHz is plotted in Fig. 12(b). The gain

pattern has improved and the magnitude of the main lobe is increased to 4.09 dBi.

Fig. 11: The magnitude and phase of the simulated reflection coefficient of the dual band antenna with the

AMC plane in proximity of the body model

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Fig. 12: The simulated antenna gain [dBi] in the vertical (xz) plane at 434 MHz (a) and at 868 MHz (b) in proximity of

the body model

V. CONCLUSION

A dual band wearable textile antenna based on substrate integrated waveguide technology that operates in the 434 MHz and the 868 MHz ISM bands, is presented. The dual band antenna was designed to have a bandwidth of 20 MHz at each resonance frequency (424 MHz to 444 MHz and 858 MHz to 878 MHz). The ease of integration into the garment of the wearer was the primary design requirement. The applied miniaturization techniques kept the dual band antenna portable while the radiation properties were still acceptable. In addition, the dual band antenna was excited by a low-complexity probe feed. An AMC plane with a zero crossing of the phase at 434 MHz, was introduced to increase the front-to-back ratio of the monopole. Simulations have shown that the AMC successfully shields the dual band antenna from the human body and that the bandwidth in both ISM bands has increased. In conclusion, the structure (dual band antenna with AMC plane) is able to successfully cover both bands of operation.

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[16] A. Vallecchi and A. G. Schuchinsky, ”Entwined Planar Spirals for Artificial Surfaces,” IEEE Antennas and Wireless Propagation Letters, vol. 9, pp. 994-997, 2010.

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List of Figures

2.1 The topology of the eighth-mode cavity antenna, excited by a low-complexity probe feed . . . 8 2.2 The simulated reflection coefficient of the eighth-mode cavity antenna in free-space 9 2.3 The simulated free-space antenna gain [dBi] in the vertical (xz) plane . . . 10 2.4 The magnitude of the measured reflection coefficient of the eighth-mode cavity

antenna in free-space . . . 11 3.1 The topology of the dual band antenna, excited by a low-complexity probe feed 13 3.2 The density of the surface current at 434 MHz . . . 14 3.3 The first implementation of the dual band antenna with a single feed . . . 15 3.4 The magnitude of the simulated reflection coefficient of the dual band antenna

in free-space . . . 16 3.5 The simulated free-space antenna gain [dBi] in the vertical (xz) plane at 434 MHz 17 3.6 The simulated free-space antenna gain [dBi] in the vertical (xz) plane at 868 MHz 17 3.7 The magnitude of the measured reflection coefficient of the dual band antenna

in free-space . . . 18 3.8 The influence of the relative permittivity on the magnitude of the simulated

reflection coefficient . . . 18 3.9 The cross-section of the body model . . . 19 3.10 The influence of the gap size between the antenna and the body on the simulated

radiation efficiency . . . 20 3.11 The influence of the gap size between the antenna and the body on the magnitude

of the simulated reflection coefficient . . . 20 3.12 The simulated antenna gain [dBi] in the vertical (xz) plane at 434 MHz with a

15 mm gap size between the dual band antenna and the body model . . . 21 3.13 The simulated antenna gain [dBi] in the vertical (xz) plane at 868 MHz with a

15 mm gap size between the dual band antenna and the body model . . . 21 3.14 The magnitude measured reflection coefficient with a gap size equal to 5 cm

between the dual band antenna and the body . . . 22 4.1 Topology and cross-section of the square patch unit cell . . . 25 4.2 The magnitude and the phase of the simulated reflection coefficient of the square

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4.3 Topology and cross-section of the loaded Koch patch unit cell . . . 27

4.4 Simulated reflection coefficient of the loaded Koch patch unit cell . . . 28

4.5 Topology and cross-section of the first version of the square patch slotted unit cell 29 4.6 Simulated reflection coefficient of the square slotted patch unit cell (version 1) . 29 4.7 Topology and cross-section of the second version of the square patch slotted unit cell . . . 30

4.8 Simulated reflection coefficient of the square slotted patch unit cell (version 2) . 30 4.9 Topology and cross-section of the I-shaped patch unit cell . . . 31

4.10 Simulated reflection coefficient of the I-shaped patch unit cell . . . 32

4.11 Topology and cross-section of the Jerusalem cross slot unit cell . . . 33

4.12 Simulated reflection coefficient of the Jerusalem cross slot unit cell . . . 33

4.13 Topology and cross-section of the spiral unit cell . . . 34

4.14 Simulated reflection coefficient of the spiral unit cell . . . 35

4.15 Topology and cross-section of the interwoven spiral unit cell . . . 36

4.16 Simulated reflection coefficient of the interwoven spiral unit cell . . . 36

5.1 The topology of the AMC surface (unit cell = 85mm x 85mm) . . . 39

5.2 The magnitude and phase of the simulated reflection coefficient of the AMC plane in free-space (unit cell = 65mm x 65mm) . . . 40

5.3 The magnitude of the simulated reflection coefficient of the dual band antenna with the AMC plane in free-space (unit cell = 65mm x 65mm) . . . 41

5.4 The magnitude of the simulated reflection coefficient of the dual band antenna with the AMC plane in free-space (unit cell = 83mm x 83mm) . . . 41

5.5 The simulated free-space gain of the structure [dBi] in the vertical (xz) plane at 434 MHz . . . 42

5.6 The simulated free-space gain of the structure [dBi] in the vertical (xz) plane at 868 MHz . . . 42

5.7 The simulated free-space gain [dBi] of the AMC plane in the vertical (xz) plane at 434 MHz . . . 43

5.8 The influence of the gap size on the magnitude of the simulated reflection co-efficient of the dual band antenna with the AMC plane in proximity of a thick layer of human skin (30 mm) . . . 44

5.9 The magnitude and phase of the simulated reflection coefficient of the dual band antenna with the AMC plane in proximity of the body model . . . 45

5.10 The simulated antenna gain [dBi] in the vertical (xz) plane at 434 MHz . . . 46

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List of Tables

3.1 Electrical parameters of the human tissues . . . 23 5.1 Radiation properties in free-space of the structure at 434 MHz and at 868 MHz 42 5.2 Radiation properties of the structure in proximity of the human body at 434

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xvi

List of abbreviations

AMC Artificial Magnetic Conductor

EIRP Equivalent Isotropic Radiated Power IoT Internet of Things

ISM Industrial, Scientific and Medical PIFA Planar Inverted-F Antenna PMC Perfect Magnetic Conductor SIW Substrate Integrated Waveguide SAR Specific Absorption Rate

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Chapter 1 Introduction

The Internet of Things (IoT) is a rapidly evolving research field and its applications form the research topic for a variety of studies. The IoT consists of a number of smart devices that are connected to a network and are able to communicate with each other. Several networks are possible: mobile networks, LoRa-networks, Ethernet-connections… The LoRa-network is specifically designed for IoT-applications. The smart devices contain several sensors to collect information of their environment and sends them over the network. At a central place, the information is collected and analyzed. It is expected by Proximus that 150 billion devices worldwide will be connected to the IoT by 2025. The number of smart devices will further increase and fulfill a more prominent role in our everyday life. Smart devices try to make our life easier and more comfortable. Their applications are uncountable and vary from a smart shower head that can play your favorite music to smart textiles. These textiles are able to integrate several electronic components such as sensors to detect body and environmental parameters or an actuator that can give a signal to the wearer. These textiles also contain a communication device that establishes a wireless communication link via an antenna to a nearby base station.

1.1 Design criteria

The antenna could be manufactured with textile materials. These textile antennas have specific design requirements and the proximity of the human body makes the design more challenging. The ease of integration into the garment of the wearer is the most important design requirement. A bulky, heavy, large antenna will not be appealing to potential customers and will therefore not be a commercial success. The ease of integration is mainly determined by the size of the antenna. Unfortunately, the size reduction comes at a prize: smaller antennas lead to a poorer performance. Besides the dimensions of the antenna, the ease of integration is also determined by the weight, the flexibility and the complexity of the feeding structure of the antenna. A second important design criterium is that the antenna must be isolated from the human body.

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2 The bandwidth should be large enough to compensate the frequency shifts, caused by the deformations of the antenna during its use or the proximity of the human body. The proximity of the human body can also cause a deformation of the radiation pattern. Therefore, it is important that the antenna has a high front-to-back ratio. The antenna will then enable more robust and energy efficient wireless communication. Additionally, less energy will be dissipated into the human body and the likelihood of exceeding the Specific Absorption Rate (SAR) limits will be smaller.

1.2 Fabrication technology

An antenna can be easily integrated into the garment of the customer when the antenna remains in the horizontal plane [14]. Therefore, Substrate Integrated Waveguide (SIW) technology is a suitable technology for the fabrication of the antenna. This kind of antennas are waveg-uide structures that contain two parallel metal/conducting planes with in between a dielectric substrate. The two parallel planes are connected by two rows of conducting cylinders or slot-s/eyelets. There are three major mechanisms that contribute to the overall loss of a SIW structure. The first major loss mechanism is due to the finite conductivity of the metal walls and are called the conductor losses. These losses can be reduced by increasing the substrate thickness. The second, predominant loss mechanism is due to the lossy dielectric material (the dielectric losses). These losses depend only on the dielectric material and not on the SIW struc-ture. The last major loss mechanism is caused by the energy leakage through the gaps (the radiation losses). These losses can be minimized/neglected if the designer places the metallic eyelets close enough to each other. The proposed design rule states that the ratio s/d (s is equal to the distance between the centers and d is the diameter of the eyelets) should be smaller than 2.5 to form a continuous wall. [14] recommends a ratio s/d equal to 2. In case the design rules are followed, are propagation characteristics of the SIW antenna similar to the classical rectangular waveguide.

The fabrication process can become very complicated due to a large number of eyelets (punc-turing steps) as well as embroidery steps. Researchers have tried to decrease the number of fabrication steps, in order to make the SIW technology more appealing for the industry. The metal eyelets can be omitted by cutting the top and bottom metal layers in one piece from a single e-textile sheet and fold them around the substrate [5]. The embroidery steps can then be avoided by making use of an adhesive sheet to attach the metal layers to the substrate. This fabrication method leads to a more light-weight, compact and mechanically more flexible an-tenna. The robustness of the antenna is increased because tearing of the electrotextile around the metal eyelets has now been avoided.

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1.3 Miniaturization techniques

1.3.1 Based on the shape of the antenna

The shape of the antenna can be exploited to reduce its dimensions. The basic idea is to use the available space more efficiently. The meander technique tries to fill the space more efficiently by bending a long straight line to occupy an overall shorter length [15]. Meander antennas are compact and easy to fabricate but can suffer from a poor gain as can be seen in Figure 1.1. This figure shows that only a part of the antenna contributes to the far field since the currents that are 180° out of phase, will cancel each other out in the far field.

Figure 1.1: Principle of a meander line antenna [15]

The fractal antenna is closely related to the meander antenna. Fractal antennas use a space-filling curve (e.g. Koch curve, Hilbert curve, Peano curve...) to construct a complex shape. The technique is an iterative process where previous segments (fractals) are broken into smaller segments, providing a greater length and so a lower resonance frequency. A different approach is to load the antenna structure with inductive/capacitive loads. These loads will create a time delay that slows down the wave propagation and the resulting antenna will look electrically longer [19]. Furthermore, the antenna is modeled as a transmission line. The researchers in [19] discovered that calculating the slow wave enhancement factor based on the loaded unit cell of the equivalent transmission line model, agrees very well with the miniaturization factor. The slow wave enhancement factor is the ratio of the loaded to the unloaded propagation constants and makes it possible to know the load parameters in the circuit model for a specific size reduction. The paper [19] proves this claim by designing two small radiators: a high-frequency slot loop antenna and a planar inverted-F antenna (PIFA). Both antennas are periodically loaded with shunt capacitors. Unfortunately, the size reduction comes at the expense of a degraded gain and bandwidth.

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4

1.3.2 Based on the substrate of the antenna

A large and bulky antenna can also be avoided when the material of the substrate and its dimensions are carefully chosen. The fringing fields of the antenna are responsible for the radiation and will concentrate where the speed of light is the smallest or where_𝜖_𝑟 is the largest. Choosing a large _𝜖_𝑟 will lead to smaller dimensions but also to smaller fringing fields and radiation efficiency [9]. The decrease of radiation efficiency could be countered by increasing the substrate height so the fields underneath the patch could easier escape. A higher substrate will extend the efficiency and the bandwidth of the antenna [4]. Increasing the height can lead in some antenna structures to the introduction of surface waves. These kind of waves travel within the substrate and are scattered at bends and surface discontinuities (e.g. truncation of dielectric and ground plane). Most of the time, they are not desirable because they dissipate power and have a negative influence on the antenna pattern and its polarization characteristics. Surface waves can be avoided by making use of cavities.

Magneto-dielectric substrates can be used to miniaturize the dimensions of the antenna [16] and is a material with both the_𝜖_𝑟 (relative permittivity) and_𝜇_𝑟 (relative permeability) greater than one. This is an interesting miniaturization technique because it leads to a smaller interaction between the antenna current and its image. Furthermore, it increases the radiation efficiency and the bandwidth. The realization techniques of magneto-dielectric substrates can be catego-rized into two different approaches. One approach places a resonating embedded circuit into the dielectric substrate. Unfortunately, this approach leads to bulky, lossy substrates with anisotropic behavior because the relative permittivity is only larger than one in the direction of the embedded circuit. A more promising approach creates the magneto-dielectric substrates through material synthesis. A large relative permittivity is obtained by compounding metal magnetic particles with dielectrics.

1.3.3 Based on the ground plane of the antenna

The antenna can be isolated from its environment by replacing the metallic ground plane with an Artificial Magnetic Conductor (AMC) plane. This structure increases the front-to-back ratio to reduce the interaction between the antenna and the human body. An AMC structure is a periodic planar array of conductive elements on a grounded dielectric substrate [26]. A realization of an AMC structure is presented in [26] and describes the design of a non-resonant co-planar monopole antenna on a miniaturized AMC structure operating in the sub-GHz frequencies. The radiating structure is realized with conductive nano-ink on a paper substrate and can therefore be fabricated with a commercial desktop inkjet printer.

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link. A metallic ground plane cannot be used to shield the dual band antenna from the human body because the electric dipole is than less efficient due to the induced anti-phase image. Replacing the electric dipole by a magnetic dipole does not result in the desired solution. A magnetic dipole is realized by a slot or an aperture. Furthermore, parallel plate modes are excited if the distance between the slot and the conducting surface is less than a quarter wavelength [10]. These modes result in a significant energy leakage and a distortion of the characteristics.

A possible solution consists of replacing the normal metallic ground plane by an AMC plane. A normal metallic ground plane has a reflection coefficient equal to -1. The current on the antenna and its image are out of phase and destructive. The reflection coefficient of an AMC plane is +1 at his resonance frequency. Therefore, the AMC plane can be placed in close proximity to the parallel current on the antenna forming a reflector, since the parallel current is in phase with its image on the AMC plane. The bandwidth of the AMC plane is defined by the phase of the reflecting wave lying in the interval 0◦ to ±90◦. The AMC plane operating in the 5 GHz band is proposed in [27]. The unit cell is constructed with a square patch because only a single resonance is required. The via connecting the center of the squared patch to the ground plane is omitted. Researchers have discovered that omitting the via facilitates the fabrication and has a negligible effect on the performance of the AMC structure for normal incidence [6]. The simple shape of the patch eases the fabrication and decreases the risk for manufacturing inaccuracies. Moreover, the square patch unit cell has a broad bandwidth [27].

1.3.4 Based on mode profile

A cavity antenna can be miniaturized based on the mode profile of the magnetic field. This technique starts from a cavity, which resonates at his fundamental mode and has 6 electric walls/perfect conductors. The tangential component of the magnetic field vanishes alongside the symmetry planes of the cavity, so they act like a Perfect Magnetic Conductor (PMC). A half-mode structure is obtained by bisecting the cavity antenna alongside the symmetry plane. The new structure has a quasi-identical mode profile and operating frequency but only occupies half of the space of the original structure. By applying this technique along the different symmetry lines of the cavity: quarter-mode [22], eight-mode [20], 16th-mode [1] or even 64th-mode antennas [23] can be created.

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6 A cavity antenna has a narrow bandwidth since only one mode is excited inside the cavity. Therefore, a miniaturization technique is often applied in combination with a technique that enhances the bandwidth. The bandwidth is proportional to the inverse of the Q-factor [9]. This factor can be lowered by increasing the height of the substrate (larger apertures) or choosing an antenna shape with more apertures through which the power can be radiated. Therefore, a circular half-mode structure will achieve a larger bandwidth than a rectangular half-mode structure [28]. Another example of an altered shape can be found in [21]. A cavity backed slot antenna is shaped to a compact half-diamond half-mode SIW on-body antenna. The design process starts from a cavity backed slot antenna and applies the magnetic-mode miniaturization technique alongside the vertical symmetry plane. The feeding point is placed in the lower half of the antenna. In case that the circular polarization requirement can be omitted, can an additional size reduction be achieved by placing a shorting pin in the upper half of the antenna. The last design step places an additional slot in the lower half of the antenna. This extra slot is used for the fine-tuning of the second order mode resonance frequency and to increase the bandwidth in the higher frequency range.

A different technique that enhances the bandwidth, is based on frequency bifurcation. Two resonators with slightly different resonance frequencies are placed close together on a common ground plane [5], [20]. The two resonance frequencies will push each other away depending on the strength of the coupling between the two resonators. The bandwidth of the resulting antenna can then be controlled by adjusting the coupling or the distance between the resonators.

1.4 Goals

A lot of research has been conducted on textile antennas. The current academic literature is mainly focused on textile antennas that operate in the GHz frequency bands and less in the sub-GHz frequency bands. This thesis outlines the design of a dual band SIW antenna, covering the 434 MHz and the 868 MHz Industrial, Scientific and Medical (ISM) band. The textile antenna is intended for an application in sub-GHz ISM bands such as those used for LoRa communication. A bandwidth of 20 MHz at each resonance frequency (424 MHz to 444 MHz and 858 MHz to 878 MHz) should be sufficient [12]. The dual band antenna is aimed to be integrated into the garment of rescue workers to provide the communication between the rescue worker and the base station. The design process of the dual band antenna is described in chapter 2 and 3. The dual band antenna has a low front-to-back ratio at 434 MHz and its performance is enhanced with an Artificial Magnetic Conductor (AMC) plane. Chapter 4 discusses different AMC unit cells and the design of the dual band antenna with the AMC plane is explained in chapter 5.

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Chapter 2 Eighth-mode Substrate Integrated

Waveguide (SIW) antenna designed for

868 MHz

2.1 The design of the eighth-mode cavity antenna in

free-space

This chapter describes the design of a textile antenna operating at 868 MHz. The Substrate Integrated Waveguide (SIW) technique is used to construct an antenna with excellent on-body performance. The design of the antenna is based on the procedure described in [20]. The procedure designs an eighth-mode SIW cavity antenna based on the virtual magnetic wall technique, that is able to meet all the design requirements mentioned in the introduction. The virtual magnetic wall technique is a miniaturization technique based on the mode profile of the magnetic field. The tangential component of the magnetic field vanishes alongside the symmetry planes of the cavity, so they act like a Perfect Magnetic Conductor (PMC). A half-mode structure is obtained by bisecting the cavity antenna along the symmetry plane. The new structure has a quasi-identical mode profile and operating frequency but only occupies half of the space of the original structure. An eighth-mode cavity antenna is constructed by applying this technique three times. The procedure from [20] enhances the bandwidth by placing two such resonators with slightly different resonance frequencies in close proximity on a common ground plane. The fields of both cavities will couple and only one cavity (main resonator) requires a feeding network while the other cavity acts as a coupled resonator. The coupled resonator is omitted to reserve space for a resonating structure operating at 434 MHz and will be added in the next chapter to construct a dual band antenna.

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8 The topology of the proposed eighth-mode cavity antenna is depicted in Figure 2.1. The dimensions of the antenna are equal to 150 mm x 150 mm x 8 mm. The eighth-mode structure radiates through its two open sides (apertures) and the tangential components of the electric field across these open sides combine in the far field. The ground plane is extended beyond the radiating apertures to obtain a more directive radiation pattern [20]. Ideally, the size of the ground plane is infinitely large to have a very directive radiation pattern and thus a high isolation between the antenna and the human body to avoid resonance frequency detuning or radiation pattern deformation. However, the ease of the integration of the antenna into the garment of the wearer is the primary design requirement and is mainly determined by the size of the antenna. Therefore, the size of the ground plane is kept limited.

The antenna is excited by a low-complexity probe feed. The connector ground is connected to the bottom of the cavity and the signal pin to the top of the cavity. The low complexity feeding structure eases the integration of the antenna into the garment of the wearer. The position of the feeding pin is chosen such that the impedance of the feed line matches the local impedance of the cavity and is optimized using CST microwave studio.

Figure 2.1: The topology of the eighth-mode cavity antenna, excited by a low-complexity probe feed

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reflection coefficient stays well below the -10 dB over its entire operational band (yellow rect-angle). The simulated antenna has a resonance frequency equal to 866 MHz and a bandwidth of 40 MHz.

Figure 2.2: The simulated reflection coefficient of the eighth-mode cavity antenna in free-space

The simulated free-space antenna gain is visualized in Figure 2.3 (red curve). This figure shows that the antenna has a good front-to-back ratio. The magnitude of the main lobe is equal to 4.17 dBi and is directed alongside the positive z-axis. The angular width of the antenna is equal to 101.1°. The green curve indicates the front-to-back ratio. The magnitude of the side lobe towards the body has a magnitude of -5.23 dBi.

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10

Figure 2.3: The simulated free-space antenna gain [dBi] in the vertical (xz) plane

2.2 Manufacturing of a prototype

The antenna must be flexible and have a good radiation efficiency. Therefore, it is important to use a substrate with low losses and an e-textile with a low sheet resistance. The radiation efficiency can be further improved by reducing the relative permittivity of the substrate but this increases its dimensions. The substrate is realized with a closed-cell expanding rubber (𝜖_𝑟

= 1.32, tan 𝛿 = 0.015) and a copper-plated taffeta polyester (𝜎 = 3 · 105 _{S/m) is chosen as}

an e-textile for the antenna. The 𝜖_𝑟 of the substrate is not exactly given by the manufacturer and can fluctuate between different points. This value is estimated by aligning the reflection coefficient of the simulation with the measured reflection coefficient of a prototype by adapting the value of _𝜖_𝑟.

The eighth-mode cavity antenna is manufactured according to the fabrication method presented in [5]. The top and bottom metal layers are cut in one piece from a single e-textile sheet and folded around the substrate. The sizing of the antenna is done by hand because the dimensions of the antenna (see Figure 2.1) are large enough to achieve a sufficient accuracy. The e-textile of the antenna is glued to the substrate by using an adhesive sheet. The substrate of 8 mm is constructed with two layers of 4 mm and an adhesive sheet is applied on both layers to glue them together.

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of the antenna is able to cover the frequency range of 858 MHz to 878 MHz. The difference between the magnitude of measured and the simulated reflection coefficient can be attributed to the inaccuracies caused by cutting the different elements by hand and the variations of the material properties such as the permittivity of the substrate.

Figure 2.4: The magnitude of the measured reflection coefficient of the eighth-mode cavity antenna in free-space

It was planned to take a clear photo of the prototype when the radiation pattern should be measured inside the anechoic chamber. The installation of the new anechoic chamber and, subsequently, the lock-down measures due to the COVID-19 epidemic (2020), obstructed the measurement of the radiation pattern.

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12

Chapter 3 Dual band Substrate Integrated

Waveguide (SIW) antenna designed for

434 MHz and 868 MHz

3.1 The design of the dual band antenna in free-space

This chapter outlines the design of a dual band antenna that operates at 434 MHz and 868 MHz. The frequency band around 868 MHz is implemented with an eighth-mode cavity antenna, which is described in the previous chapter. Unfortunately, the dimensions of an eighth-mode cavity antenna operating at 434 MHz, would be too large and thus the antenna structure would no longer be portable. This frequency band should therefore be covered by another radiating structure. In the first design, a separate feed was foreseen for each frequency band. In this first design, the frequency band around 434 MHz would be covered using a planar inverted-F antenna (PIFA). The idea of a dual band antenna was abandoned because the purpose of a dual band antenna is lost in that case. It is better to use two separate single band antennas optimized for their specific frequency region. Instead, a dual band antenna with a single feed is designed. The PIFA is feeded at the place where the local impedance is equal to 50 Ω and would be difficult to combine with the optimal feeding point of the cavity antenna. Therefore, the PIFA is replaced by a monopole antenna. Unfortunately, the PIFA should be more efficient because it is better matched to the 50 Ω feed line and therefore less power would reflect back into the source.

The topology of the designed dual band antenna is visualized in Figure 3.1. The dimensions of the antenna are equal to 140 mm x 140 mm x 8 mm. The ground plane is removed below the monopole antenna and depicted in Figure 3.1b. The monopole is placed alongside the edge of the substrate of the cavity antenna. Figure 3.2 shows that the currents flowing through the monopole, have the biggest magnitudes after the widening of the monopole and decay towards the end. The parts where the current are the largest, contribute most to the far field.

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(a) Top plane

(b) Ground plane

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14

Figure 3.2: The density of the surface current at 434 MHz

Therefore, it is important that the monopole runs directly to the zone where the ground plane is removed and that the meander technique is applied at the end. The end of the monopole is placed as far as possible from the ground plane to avoid capacitive coupling between them. Otherwise, current could flow in a loop (monopole - ground plane) which deteriorates the performance of the monopole. This is the reason why the ground plane is removed in the upper left corner (see Figure 3.1b). It improves the performance of the monopole while leaving the performance of the cavity almost untouched. The radiation efficiency of the dual band antenna should be at least 70 % so only 30 % of the injected power is transferred into heat [5].

The radiation efficiency of the monopole and the eighth-mode cavity are equal to 85,24 % and 57,58 %, respectively. The radiation efficiency of the eighth-mode cavity is lower than the proposed value, which can be increased by adjusting the feed position and his inset but at the cost of a lower radiation efficiency of the monopole. Another option for a better radiation efficiency of the eighth-mode cavity is to increase the height of the substrate but this enlarges the overall dimensions of the antenna as well and the integration of the antenna into the garment of the user will be more difficult.

The monopole does not contain any sharp edges because these would lead to a concentration of the electric field and a deterioration of the performance. The monopole will be cut out by hand from an e-textile. The width of the monopole should not be too narrow and is set to 5 mm. Otherwise, it is expected that the conducting textile will unravel and too little threads will be available to conduct the current. Laser cutting in combination with a proper coating material might be able to counter the unraveling problem.

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improving the bandwidth. In the second attempt is the same principle applied but now at the beginning of the monopole where the density of the current is the largest. The widening is equal to 36 mm (see Figure 3.1a) and improves the bandwidth of the monopole from 24 MHz to 28.5 MHz.

Figure 3.3: The first implementation of the dual band antenna with a single feed. A cone is placed at the end of the monopole to increase the bandwidth

The dual band antenna is excited by a low-complexity probe feed that is placed at the beginning of the monopole. The position of the feed is marked with a small black dot in Figure 3.1a. The monopole is matched to the eighth-mode cavity with an inset feed. The width (= 6 mm) and the length (= 30 mm) of the inset are optimized using CST microwave studio.

The magnitude of the simulated reflection coefficient is given in Figure 3.4. The dual band antenna is able to cover both operational bands (424 MHz to 444 MHz and 858 MHz to 878 MHz). The resonance frequency of the monopole is equal to 432.4 MHz and the resonance frequency of the eighth-mode cavity is equal to 869.2 MHz. The monopole has a bandwidth equal to 28.5 MHz and the bandwidth of the eighth-mode cavity is equal to 29 MHz.

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16

Figure 3.4: The magnitude of the simulated reflection coefficient of the dual band antenna in free-space

The simulated free-space antenna gain at 434 MHz is depicted in Figure 3.5. One beam of the pattern is directed towards the user and the other beam is directed away from the user. The beam that is directed away from the user has a magnitude equal to 0.68 dBi. The simulated free-space antenna gain at 868 MHz is given in Figure 3.6. The pattern is much more directive in comparison to the pattern of the monopole. The magnitude of the main lobe is equal to 1.46 dBi and has a beam width equal to 146 °.

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Figure 3.5: The simulated free-space an-tenna gain [dBi] in the vertical (xz) plane at 434 MHz

Figure 3.6: The simulated free-space an-tenna gain [dBi] in the vertical (xz) plane at 868 MHz

The prototype of the dual band antenna is constructed according to the fabrication method of the eighth-mode cavity antenna in the previous chapter. The same closed-cell expanding rubber (_𝜖_𝑟 = 1.32, tan_{𝛿 = 0.015) and copper-plated taffeta polyester (𝜎 = 3 · 10}5_{S/m) is used}

to construct the substrate and the resonator of the antenna. The pattering of the dual band antenna and its ground plane is cut by hand and folded around the substrate. The measured reflection coefficient is visualized together with the simulated reflection coefficient (dotted line) in Figure 3.7. Both measured resonance peaks are shifted to lower frequencies in comparison to the simulation. The resonance peak caused by the monopole is located at 415 MHz and the resonance peak caused by the cavity antenna is located at 854 MHz. On the figure is seen that the prototype does not entirely cover the frequency range 424 MHz to 444 MHz and that the prototype is just be able to cover the frequency range 858 MHz to 878 MHz.

Figure 3.8 visualizes the influence of the relative permittivity on the magnitude of the simulated reflection coefficient. The resonant peak of the cavity antenna shifts to lower frequencies if the value of the relative permittivity increases. The resonance peak of the monopole is relatively insensitive to variation of the relative permittivity. This is expected because the fields of the monopole are mainly concentrated in the air and not in the substrate. This figure indicates that the shift of the resonance frequency of the monopole in Figure 3.7 is mainly caused by inaccuracies during the fabrication and that the shift of the resonance frequency of the cavity is a combination of inaccuracies during the fabrication and variations of the material properties such as the permittivity of the substrate.

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18

Figure 3.7: The magnitude of the measured reflection coefficient of the dual band antenna in free-space

Figure 3.8: The influence of the relative permittivity on the magnitude of the simulated reflec-tion coefficient

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body can cause a detuning of the resonance frequency or a deformation of the radiation pattern [20]. Operating in the proximity of the human body makes the antenna design much more chal-lenging. The simulation becomes very complex when all the different influences of the human body on the antenna are taken into consideration. Therefore, it is assumed that the antenna is not bent or deformed during its use. The human body is simulated with a multilayered tissue model [30]. The cross-section of the model is presented in Figure 3.9 and contains 6 layers. The electrical parameters of each human tissue are frequency-dependent and listed in Table 3.1 at the end of this chapter [3],[24], [25].

Figure 3.9: The cross-section of the body model

The influence of the human body on the performance of the dual band antenna is investigated by varying the gap size between the antenna and the textile layer (_𝜖_𝑟 = 2.231, _{𝜎 = 0.0366} S/m) of the body model from 0 mm to 24 mm. The radiation efficiencies for the different gap sizes are presented in Figure 3.10. The radiation efficiency of the dual band antenna should be at least 70 % so only 30 % of the injected power is transferred into heat [5]. The radiation efficiency in free-space at 434 MHz and 868 MHz are equal to 85,24 % and 57,58 % (see previous section). It is immediately noticeable on the figure that the radiation efficiency is drastically reduced. The influence of the human body is more pronounced on the radiation efficiency of the monopole than on the eighth-mode cavity antenna. This is explained by the less directive radiation pattern of the monopole and a large part of the field is therefore located inside the human body. Consequently, the human body will have a larger influence on the performance of the monopole than on the performance of the eighth-mode cavity. The radiation efficiency of both frequencies improves with an increasing gap size but the radiation efficiency of the monopole remains small even if the gap size is increased to 24 mm. A poor radiation efficiency can be unacceptable for the application that is connected to the antenna. The application will be portable and likely to be battery-powered. Therefore, energy efficiency is a very important design requirement of the application.

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20

Figure 3.10: The influence of the gap size between the antenna and the body on the simulated radiation efficiency

Figure 3.11: The influence of the gap size between the antenna and the body on the magnitude of the simulated reflection coefficient

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3.12 and Figure 3.13. The gain pattern of the monopole (see Figure 3.12) is less directional in comparison to the gain pattern of the eighth-mode cavity antenna (see Figure 3.13). A large part of the fields of the monopole are radiated towards the human body such that the proximity of the human body will have a larger impact on the performance of the monopole antenna.

Figure 3.12: The simulated antenna gain [dBi] in the vertical (xz) plane at 434 MHz with a 15 mm gap size between the dual band antenna and the body model

Figure 3.13: The simulated antenna gain [dBi] in the vertical (xz) plane at 868 MHz with a 15 mm gap size between the dual band antenna and the body model

Figure 3.14 shows the magnitude of the measured reflection coefficient of the prototype in proximity of the human body. The prototype and its dimensions (140 mm x 140 mm x 8 mm) are described in the previous section of this chapter. The gap size between the antenna and the human body is equal to 5 cm. A straight low-complexity probe feed is used to excite the antenna. The SMA cable is therefore directed towards the user and unfortunately gets in the way when the antenna is attached to the chest of the human body. The difficult attachment together with the inaccuracies of the fabrication and the variations of the material parameters, result in a deviation between the measurement and the simulation.

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22

Figure 3.14: The magnitude measured reflection coefficient with a gap size equal to 5 cm between the dual band antenna and the body