A DC to 3-phase series-resonant converter with low harmonic distortion

(1)

A DC to 3-phase series-resonant converter with low harmonic

distortion

Citation for published version (APA):

Huisman, H., & Haan, de, S. W. H. (1985). A DC to 3-phase series-resonant converter with low harmonic distortion. IEEE Transactions on Industrial Electronics, 32(2), 142-149. https://doi.org/10.1109/TIE.1985.350185

DOI:

10.1109/TIE.1985.350185

Document status and date: Published: 01/01/1985 Document Version:

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(2)

A DC

to

3-Phase

Series-Resonant

Converter with

Low

Harmonic

Distortion

H. HUISMANANDS. W. H. DE HAAN

Abstract-A type of dc to 3-phase series-resonant converter (s.r.-converter) for potentially submegawatt industrial applications is pre-sented. The converter provides variable-frequency sine-wave currents, with low harmonic distortion at the output terminals, and with the frequency ranging from-200 throughdc to +200 Hz.The converter can transfer power in both forward and reverse power-flow directions to almostany typeof load circuit. Themethods of controlareformulated suchthattheycanbeimplemented easily with high-speed logical circuits. Testresults fora1-kW demonstration converter aresupplied.

I. INTRODUCTION

DURING the last decadethe interest in electrical

wind-en-ergy conversion has increased significantly. Thisattention concerns both the generation of autonomous multiphase ac

grids,as well as the delivery of power toexistingutilitygrids.

To carryoutthe electricalconversionprocess,oneoftenuses a

cycloconverter or a cascaded ac-dc and dc-ac converter. The dc to3-phaseconverterpresentedinthispaper isconsideredas an intermediate step in the development of a novel type of 3-phase-to 3-phasecycloconverter.Apart fromwind-energy con-version this converter canbe appliedtootherareasof

technol-ogyincludingpowergenerationandvariablefrequency-machine driving.

Fundamentals of dc to 3-phase ac power conversion have been known for many decades. Because of the simplicity of control and the clarity of insight into the method of opera-tion, some of these systems, such as the load-commutated bridge inverter, the square-wave inverter, and the

high-fre-quency pulsewidth-modulated inverter,arewelldeveloped [1]

andwidelyspread. The advantages of these systemsare,

how-ever,countered byoneor more of thefollowing disadvantages:

1)the stress on electronic switches is high due to forced

commutationtechniques,

2)heavy and costly filters are required to remove the

low-frequency harmoniccontentof the electricalwaveforms,and

3) thepowerfactorispoor and cannotbe controlledfreely. The s.r.dcto3-phaseacconverterpresented,whichisbased

on aconceptpresented by Schwarz [2],doesnotpossessthese

disadvantages. Moreover, the system has thecommonfeatures of series-resonant converters (s.r.-converters) such as, 1) the system isinherently short-circuit proof, 2) the sizeof the

sys-tem is smallduetothehigh internal frequency,and 3) the

sys-tem is capable ofcontrolling and turning off power bya

frac-tion ofa millisecond. The technology of the single-quadrant

Manuscript receivedSeptember, 1984.

The authorsare with the DelftUniversity ofTechnology, Mekelweg4,

2624,CDDelft,TheNetherlands.

ipt

pPI1PIIPIPPII11111 I I I IiIiiil I i ilIIIlI I I P

TTP

t-iqt I ili I I II h Ii i liifi

irt~~P1~~~~PPbP1i||1111|l1PP 'I'i UFPI IF IGwF1w|1

Fig. 1. Example ofthe distributionofthe currentpulses in theresonant

currenti1overthe line currentsip, iq,andi4.

dc-dc s.r.-converter has been well established by now and

manyauthors have discussed thesubject [3] -[6].

Sincethe beginning of theeighties attemptshave been made

to breakthroughtothedevelopment ofacandmultiphase s.r.-converters. Some successful experiments of ac-dc

s.r.-con-verters were reported by Schwarz [7], [8].Recently, a single-phase dc-ac converter has been presented [9]. The

30-dc

(3-phase to dc) s.r.-converters described up to now [7],

[8]

do, however, extract currents from the power grid which are

similar to the currentswhich areextracted by a3-phasediode

rectifier. These square-wave-like currents cause harmonics in the supply lines, while thepowerfactoristheoretically limited

to0.955 [8].

From a fundamental point of view, the s.r.-converter is

capableofdealing withmultiphaseacpowerina more sophisti-cated way [2].A s.r.-converter canbe consideredas anetwork

whichgenerates an array ofcurrentpulsesat ahighfrequency

(5-50 kHz). Byproperly distributing the pulses from this high-frequency carrier over the 3 lines of the grid, it is possible to generate current waveforms, which do contain one single fre-quency in the low-frequency range (<3 kHz).Theprinciple of the process is illustrated in Fig. 1, where resonant current

pulses are represented by impulses. Fig. 1 is obtained from a

simulation program, which operates similarly to the one

de-scribedin

[10]

.Thehigh-frequency contentofthe output

cur-rent is removed by a relatively small filter, with the result

being a low-frequency phase current with low harmonic

dis-tortion. The pulse distribution process can be applied bothto

30-dc

convertersaswell asto

dc-34

converters.

Inthis paper a dc-30converter is described whichisbased

on the above principle. The converter is capableof generating

a 3-phase output waveform with frequencies rangingfrom ca

+200 via 0to-200 Hz(phase reversal).Moreover, the conver-ter can beconnectedtoleadingaswellaslaggingloads(+

1800

to -180°),both linear and nonlinear, inboth the forward and

i1

t I 11i tI I I I i I I i i I iI k i I I I 11i Ii I I I II I I I iI I i I

(3)

reverse power-flow directions. The critical points thatcan be identifiedin the system design of the dc-34 converter are: 1) the pulse-distribution algorithm (line selection), 2) the

algo-rithm for the selection of particular switches in the selected lines (switch selection, and 3) the method of control to guarantee thyristor turnoff and continued oscillation (Vs-control). The method of controlis in accordance with a

con-trolmode, which isdescribed in acompanion paper [11].The

theory is verified ona small-scale (1 kW) demonstration con-verter.

II.PROPERTIES OFTHEGENERALIZED DC-DC

S.R.-CONVERTER

The operationmodesofthe

dc-30

s.r.-converter isbased on

the principles of the dc-dc s.r.-converter, which is treated in

literature [3]-[6],

[11]

, [12]. Toenable greaterinsight, some

properties of the dc-dc s.r.-converter will be reconsideredina

nonorthodox manner. Fig. 2 shows the basic, generalized dc-dc s.r.-converter. The input switch matrix consisting of

switches QA and QB connects the resonant circuit to

+E,

or

-E5,

where E is the dc source voltage. The output matrix consisting of switches

QC

and QD connects the resonant

cir-cuit to +V0 or -VO. By appropriate control of the switch

matrices a bipolar voltage, compounded of square waves, is

generated across the resonant circuit, thus inducing a

reso-nant current of thewell-knownshape(Fig.3(a)). In thispaper the resonant current is consideredas a seriesofcurrentpulses. The time ofinitiation of theKth current pulse is denoted by

ak, where : is the normalized time defined

by

:=

t/<LI

C1.

The current gap which is not present in most dc-dc

con-verters is a prerequisite in ac-converters for the purpose of

commutation and control

[11]

.Dependingontheapplication,

the switches are implemented as single diodes or as combina-tions of diodes and/or controllable switches suchasthyristors

andMOSFET's.Inmultiphaseconvertersallsymbolicalswitches shouldbe implemented asthyristors. Althoughthemultiphase

converter does not contain diodes, we will stick to the term

"diode current" foreach firstcurrentsegmentfollowinga cur-rent zero crossing; the second current segment will be

desig-nated as "thyristor current". TheKth event betweenthe time

fk and fk±1 willalso be referred toastheKthconverter

cycle.

It is well known that a number ofswitching modes canbe

distinguished for the generalized s.r. dc-dcconverter

[4],

[11].

It will be shown that all switching modes have the following

properties in common:

1) the diode current is (at least) opposed by the highest

terminal voltage (this follows from the commutation

condition),

2) the thyristor current at the high-voltage side extracts

power from that source (this follows from the power

balance),and

3) both diode and thyristor currents on the low-voltage

sidetransferpowerinthe desired direction.

Conversely, one may state that if the above rules are obeyed,

one can transfer power in any desired direction in any

com-binationofinputandoutputvoltages.

The statements are based on the

following

considerations.

EsQ4

j

j)Vo

aB

Wl04A

Fig.2. Basic schematic and thedefinitionof thevoltagesandcurrents ina

generalizeds.r.dc-dcconverter.

Suppose that in thesymbolic converter from Fig. 2, the source

voltage Es and the outputvoltage V0 are both allowed to be

positive or negative. In either an input or an output line the

current will be prescribed by some reference. The desired

power direction is defined by the sign of thisreference cur-rent and the sign of the actual voltage at the associated port. As stated before, a resonant current phase consists of a diode

current and a thyristor current. QD andQT aredefinedasthe charge transferred bythe diode and the thyristor current,

re-spectively. Both QD and QT are defined such that they are

always positive. The input and output matrices facilitate QD

and QT being routed in any desirable directionon input and

output terminals.

If one bears in mind that the energy transferred by a cur-rent in a certain interval is equal to theproduct of voltage and transferred charge, it follows from considering theenergy bal-ance applied to the input and output port ofthe converter

(Fig. 2)

ES(QDSll

+

QTS1 2)

=

VO(QDS21

+

QTS22)-

(1)

Here

SI1

to S22 indicate in which direction the charge flows

on input and output, so

Si1

iseither +1 or -1. Note that (1)

is valid under nonsteady-state conditions, provided that

v,p(k)

=

vcp(k

+ 1) and that both E and V0 are constant duringthecycle considered.

From(1)itfollowsthat

QD

S¾2Es-S22Vo

QT

S11Es

-

S21 Vo

(2)

Because both QD and QT are positive, the sign ofthe left-hand term is negative. To satisfy (2), the coefficients

Si,

and Si2 of the highest voltage, either Es or VO, should be opposite in sign. On the low-voltage side both

Sji

and Sj2 are

unconstrained, so that they are chosen such that the power is transferred in the desired direction. For a proper commuta-tion from the diode to the thyristorcurrentthe diode current is required to be opposed by the highest voltage. By way of

example Fig. 3(b) shows the resulting current waveforms of

isandio,if

Es

>0,

Vo

> 0,

Es

<

Vo

andioref < 0. The elaboration of the switch selection mechanism in the

dc-30 converter presented is primarily based on the previous discussion. The basic diagram of the power network of the

converter is depicted in Fig. 4. The switches symbolize ap-propriate electronic switches such as antiparallel thyristors.

(4)

(a)

is!

Pkf:

T

(b)

Fig. 3. (a) Resonantcurrentwaveform forconverterswithacurrent gap.(b) ExampleofsourceandloadcurrentsinaconverterwhenEs>0,V0 >0,

IEst < IV,1 andi,.f < 0.

vc

Fig.4. Basicschematicofthepowernetwork ofadc-3 s.r.-converter.

In essence, the multiphase-converter operation is based ona

selection mechanism which, for any converter cycle, selects twoout of three output linestocarrycurrents. Theparticular implementation of this line-selection mechanism will bU dis-cussed later. For our present considerations it is importart to note that once one ofthe three output lines has been

ex-cluded from current flow, during this particular converter

cycle, the converter can be treated as a generalized dc-dc converter.

A.BasicRequirements

Prior to describing the control and selection mechanisms,

we formulaterequirementswhich the ac-converterhastomeet. Theconverter should be abletoinject currents inanyarbitrary

active or passive output circuit. These currents will often be

sinusoidal ofshape. The size and shape ofthe voltages at the output terminals depend on the character ofthe load and the

shape of the injected currents. The amplitudes of the output

voltages are allowed to be greater as well as smaller than the source voltage. For that reason the converter should be able to transfer power to a lower as well as to a higherterminal

voltage. Moreover, because the load may require reactive

power, the instantaneouspower should be allowed to flow in both directions. The output capacitors of the converter are

sufficiently large

to

justify

the assumption that the

output

voltage

is constant

during

a converter cycle, althoughthe

out-put voltage is essentially a low frequency ac-voltage. One should bear in mind that the output capacitors are intended

to remove the high-frequency content (order 10 kHz) from theoutputwaveform.

Concluding,

it is evident that

during

each converter

cycle

the

multiphase

converter should be able to operate as a gen-eralizeddc-dcconverter.

B. The

Principal

Functions and Block

Diagram

To operate as a dc-dc converter during a converter

cycle,

it is necessary to select for each convertercycle two lines on

the ac side. Two out of M lines have to be selected for an

M-phase

converter. This line-selection process and the associ-ated pulse-distribution process is controlled by the actual measured output currents im(m = p, q, r) and by the

refer-ences for the output current imref. The selection

circuit,

which is of theintegral-pulse-controller type [11],

[13

[14]

is symbolicallyindicated in the lower left of the blockdiagram

in

Fig.

5. The line-selection block also generates a signal p

whenever a current pulse should be initiated to bringone of theoutputwaveforms in accordance with the reference.

After twogridlineshave been selected, it is theknownby

measurements, to which actual input and outputvoltagethe

dc-converter will be connected. The appropriate information

is passed via a multiplexer to the switch-selection logic. The

switch-selection logic, in the middle of Fig. 5, determines

from the signs of the passed signal and from the state ofthe

converter (diode cycle, thyristor cycle,orgap)which switches have to be closed during the following converter subcycle.

Four out of20switches have tobe selected.Thisinformation

is passed to the latched thyristor pulse generators whenevera p-signal is issued during a current gap or when a cs (change

state) signalisissuedbytheso-called VJ-peakpredictor. Essentially, thep-signal initiatesthe turnover from the cur-rent gap to the diode current, while the cs-signal initiates the

turnover from the diode currenttothethyristor current. The

vup-predictor

generates a cs-signal whenever this cir-cuit predicts that the resonant capacitor voltage will be at a

specified reference levelatthe end of the convertercycle. C. Switch-Mode Selection

Once a pair of terminalshas been selected, an appropriate andunique setofswitchescanbedesignated.

Explicitly, the switch-selection process for the diode and the thyristor cycle, respectively, will be explained with

refer-ence toFig.2.

1) Diode Cycle: Based on the sign of thecapacitor voltage

vC at time

Ok,

it is known in which direction the resonant cur-rent will flow during the following converter cycle. As stated

before, the diode current should at least be opposed by the highest terminal voltage (Es or VO)- So, if one measuresE,

V,

and vu, one can straightforwardly designate two switches

on the "high-voltage side" which have to be closed. On the

(5)

then sign

(i,)

=-sign(VO)

and sign(i5)=sign(. ,)

sign(Es) (6)

Fig. 5. Blockdiagramof the controller for the dc-34 s.r.-converter. The PIC and the

Vp-predictor

doprocess analogsignals. All other circuits process booleans.

desired power transfer direction, defined byIoref Uo,is

satis-fied.

So,if

I

VO

I

<Es

then sign

(i,)

=sign(ioref)

andsign

(is)

=-sign(Es)

andif

I

VO

I>

I

then sign

(io)

=sign

(VO)

If the signsofE, VC,

'oref,

andv, are determined, the ap-propriate switches can bedesignated easily from (3) to (6). D. Line Selection

In the following section it isassumed that a s.r.-converter should feed an M-phase network with ac power. For the pur-pose of driving machinery, it is suitable if the converter be-haves like a multiphase current source, although for multi-phase power generation avoltage-source character is more ap-propriate.The line-selection mechanismwill be explained with respect to a current-source character. The multiphase

con-verter should be controlled such that the current im in line

m (m =

1,

2, **, M) is a replica ofa reference current imref

(m = 1, , M). The reference current in lines 1, ,M-1

can be chosen freely; the reference current in line Mfollows from

m

Y,

imre6f-°

m=l

(7)

The current waveform synthesis is based on the concept of pulse-integral control [11],

[121.

In a dc-dc converter the resonant current

il

is measured, subsequently rectified and compared to areference iref. The error

signal(li1

-i

i'f)

is fed to an integrator. Whenever the integrator output

ec

ex-ceeds some threshold eT, the power switches are actuated, thus generating another current pulse. When two successsive

threshold crossings are denoted by ta and tb, the following

holds: sign

(ioref

VO)

andsign

(is)

= sign (E

s)

sign(Es)

If the directions of

is

andio, areknown,and if the direction of

il

is

known,

a unique set of four

appropriate thyristors

canbe

designated. Note that sign

(ij)

in the Kth cycle follows from

sign(vc(f3k)).

2)Thyristor Cycle: The current direction on the

low-volt-age side occurs during the

thyristor

cycle

such that the

de-sired

power-transfer

direction is satisfied. On the

high-volt-age side the direction of the

thyristor

currentis

opposite

to

that of the diodecurrentonthat side. So, if

VO

<

EsI

then sign

(io)

=sign(ioref)

andsign

(is)

=sign

(Es)

(5)

and, if

VO

I

>

I

EsI

Itb

(4)

(io-

iref)dt=0.

ta

(8)

For dc and single-phase converters,ta and tb coincide with the beginningoftwosuccessive convertercycles.

For normal unipolar dc-dc s.r.-converters the

implementa-tion of the pulse-integral controller

(PIC)

for the output

cur-rent io is relatively simple because

i4

= lii In multiphase

converters only twolines are served during a converter cycle.

For that reason (8) is implemented such that the interval

(ta,

tb) effectively spans several cycles. This implies that during

each cycle alimited error occurs for each line.Moreover, the implementation should be modified such that the PIC can

handle both positiveandnegative

valuies

of

il

andiref.Fig.6(a)

gives the implementation of one of the M-1 PIC's and

Fig.

6(b) shows the associated waveforms. Whenever theintegrator output

e.

gets into one of the arced areas, a pulse-request

signalPm isgenerated.

The gap between the comparator levels is intended to pre-vent the immediate generation of a pulse request whenever

the reference signal changes

polarity.

IfA=0,thisundesirable

(6)

(a)

imreff

ect

lint

(b)

Fig. 6. BipolarPICforline m. (a)Schematic diagram.(b)Signals that may occuratthetimethat the reference signali7Jrefisreversingpolarity. Note thatattimet4 apulserequest that is issued by another line is served.

be slightly larger than the integrator swing during a current

pulse, although this is not critical. Note that signal P only

in-dicates that line m requests a currentpulse. It does not

indi-cate inwhich direction the currentpulse should flow intothe line.

All lines are equipped with individual PIC's and the

pulse-request signals are logically OR'ed. When a pulse request is

issued, the next converter cycle will be generated. The line that issues the request willtake part inthe oscillationandthe

current will flow inthat line inthe directionof theassociated reference. Next asecond line that will take partinthe

oscilla-tion should be selected. Only those lines whose sign of the

reference current isopposite the reference signofthelinethat

issues the pulse request are considered. Ifmore than one line

satisfies this requirement, a line is selected whose integrator

output is closest to the associated threshold level. Note that the line selectionprocess described above is based fora great part on two-state logic, whichcanbeimplemented easilywith

digitalelectronics.

After two lines on the ac side have been selected, the con-verter can further be considered as a generalized dc-dc con-verter. For this specific dc-dc converter, the voltagebetween the selectedlinewillbedesignatedasoutputvoltage

V',

while the output current inthe linethat issuedthepulserequest will bedenotedby

i,

The further selection ofswitchesiscarriedout asdescribed

intheprecedingsection.

During the servicing ofa pulse-request, i.e., thegeneration

of a current pulse, the pulse-request signal will inherently be cleared duetothe integratoraction.

E.

VJ-Control

One of the most critical pointsin multiphase convertersis

the choice of the firing-angle control module. This control

module should be chosen such that thyristor turn-off

condi-tionsand uninterrupted converter operation areguaranteed.

In a companion paper [11],acontrol module hasbeen

pre-sented which guarantees continuous oscillation, in particular

under dynamic conditions and at abrupt changing source and load voltages as occurring in multiphase converters. This

con-trol module maintains the capacitor peak voltage at a prede-scribed level

Vcpref,

whichshouldbeatleasttwicethehighest

terminal voltage.

Duringthe diode interval following

1k,

the circuit

continu-ously generates a real-time prediction of

vup(k)

accordingto

thefollowing formula [11]:

vcp(k)=U1 - U2 + {(U1 -

U2-VC())2

+

(Zlil)2

}11

2

(9)

where

U1

and U2 correspond to either

+E,

or

-E,

and

+±V

or -VO, depending on the current flow direction through the

sourceand loadintheinterval considered.Atatime

(,

the

pre-diction wouldbevalidif thecircuit were to turn overfrom the diode to the thyristor cycleatthat time.So wheneverthe

pre-dicted value crosses thereference value Vcpref, the turnover is

actually initiated. The beginning ofaconverter cycle at time

(k isinitiated by thelogically oR'edpulse requestsignals. Asdiscussedin [ 1],thecontrol module providesa method of independent control of the average current and capacitor

peak voltage. Note that the time betweenthe terminationof a

thyristorcycleandtheinitiationofthe nextdiodecycle should be larger than the thyristor turnoff time tq. Protection cir-cuitry is providedto delay the initiation ofa diode cycle ifa

pulse-request signal is issued before the associated thyristors

haveturned off.

Because the charge which is transferred by the resonant circuit ina converter cycle islimited to 2

vcpl,C

thisdoes in

fact signify that theconverterinherently hasacurrentlimiting character.

III.

VERIFICATION

The proposed algorithmsandtheassociated electronicswere

tested on a 1-kW demonstration converter whichhad the fol-lowingspecifications (referto Fig.4):

sourcevoltage

maximuminverterfrequency

peak capacitorvoltage

maximumallowable output voltage resonantcapacitor

resonant inductor input filter capacitor

output filter capacitors

power at full load

Es

=120V dc

fi

= 10kHz Vcp =350V

Vomax=

170V

C1

= 1.11 pF L1 = 146,uH

Cs

=

50MF

CO

= 50

,F

PO

= 1 kW

The electronic switches are implemented with 10 pairs of

(7)

Fig. 7. Pulse-distributionprocess at a lowoutput frequency. Uppertrace:

resonant currentca, 40 A/div. Lower traces: unfilteredlinecurrents, 50 A/div, 100 ps/div.

The converter was primarily intended to demonstrate the

most important features of the

dc-30

s.r.-converter such as

harmonic distortion, power factor regulation and load

inde-pendence. Optimizing the performance with respect to

ef-ficiency and size was not a goal,

although

a maximum

ef-ficiency of 85 percent was established. The relatively high

losses are due to the low source voltage in combination with the oversized power network.

Fig. 7 shows the resonant current and the unfiltered output currents at a relatively low linefrequency. Note the difference

in shape of the current pulse inthelinesp andq, whichis due

tothe fact that Vpr< Eswhile Vqr>

E,

Fig. 8 gives an impression of the pulse-distribution

mecha-nism. Simultaneously, it isshown that the capacitor peak

volt-age ismaintained within 10 percent at a constantlevel.

The filtered and unfiltered currents in line p are shown in

Fig. 9. The frequency of the reference current isapproximately 60 Hz. From the

frequency

spectrum ofFig.

10(a)

it can be concluded that theharmonic distortion of the filteredline cur-rent is extremely low. The 3rd, 5th, and 7thharmonicsare at

least 40 dB below the 100-Hz fundamental component. For comparative purposes, the harmonics of a line-commutated

converter are indicated in the same graph. The difference is

self-explanatory.

As stated in relation to (7), two reference currents can be chosen independently. For the purpose of examining the cross

distortion between lines, a frequency spectrum (Fig. 10(b))

was recordedforaconditioninwhich line p generated a 100-Hz

sinoid, and line q a 130-Hz sinoidal current. The cross

distor-tion is approximately -25 dB. Notethat the 3rd, 5th, and 7th harmonics have disappeared from the graph. The "grass" at -60 dB between the harmonics is due to the more or less stochastic character of the unfiltered line current, which is

rooted in the fact that the inverter cycle is not synchronized withthe ac-cycle. From the lastgraphitmay beconcluded that the converter presented canbeconsideredas a

multiphase

cur-rent amplifier. Any arbitrary current

iref

can be reproduced

in theoutputline.

To investigate the ability of the converter to feed currents

in nonlinear and active loads, two test circuits were set up. In the firstcircuit theconverterwasconnectedto anasynchronous machine. The converter was able to accelerate and decelerate

the machine in both rotation directions. When the axis of the machine was connected to a mechanical drive, it was possible

Fig. 8. Sinusoidal (200 Hz) modulated output currents. Upper traces: unfiltered line currents 50 A/div. Lowertrace: resonant capacitor voltage ca, 700V/div.

Fig. 9 Sinusoidal (60 Hz) modulated currents in line p. Upper trace: unfilteredcurrentinlinep, 50 A/div. Middle trace:filtered current in line

p, 10A/div. Lowertrace: reference for line p, 2ms/div.

to use the machine as a generator, thus transferring power to

the dc "source"

E,

The reactive power for the machine was inboth casessuppliedby theconverter.

In the second-setup theconverter wasconnected via a

trans-former to the 3-phaseutility grid. In order to test the ability of the converter to supply current independent of the actual output voltage, a 33 1/3-Hz

30

reference was phase lockedto

the 50 Hz grid, forcing the converter to supply a 33 1/3-Hz current. The filtered output currents, shown in Fig. 11, arise

from a superposition of the 33 1/3-Hz converter current anda

50-Hzcurrent, which iscaused by the connection of the con-verter filtercapacitors to the grid.

IV.CONCLUSIONS

A

dc-30

s.r.-converter is presented which has a combination offavorable properties comparedto other types ofmultiphase

converters.

Thesepropertiesinclude:

low sizeand weight ofthe converter;

lowharmonic distortion ofthe generatedcurrents;

ability to generate low-frequency ac currents(up to several hundredsofHz) of any arbitrary shape;

abilityto feedcurrents to almostany typeofloadincluding existingutility grids,machinery and passive loads;

potential abilitytoprocesshundreds of kW of power with an

efficiency ofmorethan 90 percent;

high speed of reaction;

inherently short-circuit proof; and bothforward and reverse power flows.

(8)

_ 3-phase AC 105 Hz spectrumof i p 1 - 1 = p q pq- qr r = 12.5 A r u = 75 V rp fifth harmonic of a LCC

---

--_I

- -

iffiLIILIIIIL_-100 200 seventh harmonic of a LCC 300 400 500 600 700 800 900 Hz (a) OdE - - - -.

~~~~~~~~~~~~~~~~spectrum

of dB

dd

t~~~~~~~~~~~~~~~~~~~

A;

100qHz

S -10 5 A; 130 Hz -20 -30 -40 --f---- -. --80 -100 200 300 400 500 600 700 800 900 (b)

Fig. 10 Frequency spectra of the line currents. (a) Spectrum of i,; the reference for linepisa 100-Hz sinoid. (b)Spectrum ofi,;thereferences for thelines p and qare100- and 130-Hz sinoids, respectively.

Fig. 11 Filtered load currents in lines p and q when the converter is

connectedtoa50-Hz3-phasegrid andwhilethe current referenceis set to 33 1/3 Hz. Uppertrace: i, , 10 A/div. Middle trace: Vpq, 100 V/div.

Lowertrace: if, 10A/div; 10 ms/div.

a dB -10 -20 -30 -40 -50 -60 -70 -80 0 F 0 Hz

(9)

The properties of the converter are rooted in

combining

a well-known s.r. power network

topology

withanovel method of preserving the energycontent of theconverter

(vp

-control),

and ofdistributing currentpulsesovertheoutput lines.

REFERENCES

[1] J. M. D. Murphy, ThyristorcontrolofACmotors. New York: Pergamon, 1973.

[2] F. C. Schwarz, "A doublesided cycloconverter,"IEEE Trans. Ind. Electron. Instrum.,vol. IECI-28,pp.282-291,Nov. 1981. [3] ,"Animprovedmethod ofresonant currentpulsemodulation for

powerconverters," IEEE Trans. Ind. Electron. ControlInstrum., vol.IECI-23, pp. 133-141, May1976.

[4] F. C. Schwarz and J. B. Klaassens, "Acontrollable 45-kW current sourcefor DCmachines,"IEEETrans.Ind.Appl.,vol.IA-15,pp. 437-444,July/Aug. 1979.

[5] R. J.Kingand T.A.Stuart, "Modellingthefull-bridgeseries-resonant converter," IEEE Trans. Aerosp. Electron. Syst., vol. AES-18, pp.449-459,July 1982.

[6] V. Vorperian and S. Cuck, "Acomplete DC analysisof the series-resonantconverter,"presentedattheIEEEPowerElectronics Special-istsConf.,(Cambridge, MA),June1982.

[71 F. C. Schwarz and W. L. Moize de Chateleux, "A multikilowatt

polyphaseAC/DCconverterwithreversible power flow andwithout

passivelowfrequencyfilters,"presented atthe10thIEEEElectronics

SpecialistsConf. (SanDiego,CA), June1979.

[8] F. C. Schwarz and J. B. Klaassens, "A controllable secondary

multikilowatt DCcurrent source with maximum power factor in its three phase supply line," IEEE Trans. Ind. Electron. Control

Instrum. vol.IECI23,pp. 142-150, 1976.

[9] J.B.Klaassens,"DCtoAC series-resonant converter system withhigh internalfrequencygeneratingsynthesised waveforms formultikilowatt powerlevels," presented at the IEEEPower Electronics Specialists

Conf., (Gaithersburg, MD),June1984.

[10] R.J.Kingand T. A.Stuart,"Alarge-signaldynamic simulation for the series-resonant converter," IEEE Trans. Aerosp. Electron. Syst.,

vol.AES-19,pp. 820-829, Nov. 1983.

[11] S. W. H. de Haan and H. Huisman, "Novel operation and control modes forseries-resonant converters," IEEE Trans. Ind. Electron.,

pp. 150-157,thisissue.

[12] F. C. Schwarz and J. B. Klaassens, "A reversible smooth current sourcewith momentaryinternalresponse fornon-dissipative control of multikilowattDC-machines,"presentedatthe IEEE SummerMeeting PowerConf., (MN), 1980.

[131 F. C. Schwarz, "Engineering information on an analog signal to discrete time intervalconverter,"NASACR-134544, 1973.

[141 S. W. H. de Haan, "A new integral pulse module for the

series-resonantconverter,"IEEE Trans. Id.Electron., vol. IE-8, pp.